The high speed and long range of modern airborne vehicles places increasing range demands on radar systems used for tracking. The long-range requirement also requires the use of relatively high transmitted power to reliably detect small targets. High transmitted power implies a relatively higher peak transmitter power or a longer duration transmitter pulse (also known as a "wider" pulse). Assuming a maximum available peak power, longer range implies a longer duration transmitted pulse. A longer duration pulse tends to reduce range resolution, which is the ability to distinguish among targets which are at similar ranges. Pulse compression techniques can be used to improve range resolution in spite of the longer pulse duration, thus eliminating the need for high peak power short pulses, but the minimum range at which a target can be detected by a monostatic radar system increases with the transmitted pulse length. Thus, if the transmitter pulse duration is 100 microseconds (.mu.s), the minimum distance at which a target may be detected by a monostatic radar is about 8 nautical miles (nm). Clearly, a radar using pulses of such a duration cannot be used to detect close-in targets, as for example aircraft which are landing or taking off from an airport at which the radar is situated. An additional problem associated with pulse compression is the appearance of range sidelobes (as distinguished from antenna sidelobes) in addition to the main range lobe. The time position, or range, of the main lobe is the position that is tested for the presence of a target and for estimating the parameters of that target (reflected energy or power, fluctuations in echo power, and closing speed, etc.). The presence of range sidelobes on the compressed pulse results in interfering echoes which originate at ranges other than the range of the main lobe. This interference, known as "flooding," can cause erroneous estimates of the echo characteristics in the range cell (i.e., range increment) covered by the main lobe. Prior art techniques for suppressing range sidelobes include the "zero-Doppler" technique, in which the range sidelobe suppression scheme is based in part upon the assumption that the interfering echoes, as well as the desired echo, are associated with a closing velocity which results in no significant Doppler phase change or shift over the duration of the uncompressed pulse. The Doppler phase shift .phi..sub.DV across the uncompressed pulse is 2.pi. times the product of the Doppler frequency shift and the uncompressed pulse duration (i.e. .phi..sub.DV =2.pi. f.sub.d T.sub.0 radians). When this Doppler phase shift is actually zero or very small, moderate sidelobe suppression is achievable with the zero Doppler design. However, the zero Doppler design is very sensitive to small Doppler frequency shifts, making deep sidelobe suppression impossible for radar applications in which such deep sidelobe suppression is desired, as for example in weather mapping, clear air turbulence detection, and microburst detection.
An approach to range sidelobe suppression elimination is described as prior art in U.S. Pat. No. 5,151,702, issued Sep. 29, 1992, in the name of Urkowitz (Urkowitz '702), incorporated herein by reference. FIG. 1 is a simplified block diagram of prior art as described in Urkowitz '702. In FIG. 1, a complex received signal I+jQ is applied by way of an input port 10 to a pulse compressor, illustrated as a block 12, for enhancing signal-to-noise ratio. The compressed signal is applied to a range sidelobe suppressor 14, which may be implemented as a mismatch filter for reducing range sidelobes which result from pulse compression. The pulse compressed, range sidelobe suppressed, echo signal is applied to a pulse-to-pulse Doppler filter bank 16, which separates the received signals from sequential receptions into frequency bins, as well known in the art. The Doppler filtered signals from Doppler filter bank 16 are independently applied to amplitude detectors 218a, 218b, 218c, . . . , 218m, for generating the desired radial velocity information of both point and diffuse targets, which may then be applied for further processing and display. The further processing may include, as indicated in FIG. 1, threshold processing for determining the presence of a target in noise and clutter. This prior art arrangement for suppressing sidelobes includes the "zero Doppler" technique, in which the assumption is made that the interfering echoes as well as the desired echo have a closing velocity that has no significant Doppler phase change or shift over the duration of the uncompressed pulse, as described in detail in U.S. Pat. No. 5,173,706, issued Dec. 22, 1992 in the name of Urkowitz (Urkowitz '706), incorporated herein by reference. As mentioned, when the Doppler phase shift is actually zero or very small, moderate sidelobe suppression is achievable with the zero Doppler design, but the design is very sensitive to small Doppler frequency shifts, thereby making deep sidelobe suppressio impossible in the presence of such shifts.
The solution to the abovementioned problems as described in Urkowitz '706 is illustrated generally in FIG. 2. FIG. 2 is a simplified block diagram of an embodiment of the invention which is better suited to larger Doppler frequency shifts and/or larger duration-bandwidth products than the structure of FIG. 1. In FIG. 2, the I+jQ signal, representing the complex envelope of the radar echo, plus whatever receiver noise is combined with the echo, is applied by way of port 210 to Doppler filter bank 216, without being pulse-compressed. Filter bank 216 separates the signal into frequency bins, and applies the signal in-each bin to a separate processor 228, which performs the functions of both pulse compression and range sidelobe suppression (PC & SS). As with the arrangement of FIG. 1, the output from the lowest-frequency bin of Doppler filter bank 216, namely the f.sub.1 bin, is applied directly to its associated processor 228a, without a multiplication or frequency conversion. The output signals from filter elements f.sub.2 though f.sub.m of Doppler filter bank 216 are individually applied to a corresponding multiplier 220. For example, the output port of filter element f.sub.3 of filter bank 216 is applied to an input of a multiplier 220c. Multiplier 220c also receives from a source (not illustrated in FIG. 2) an oscillation signal exp(-j2.pi.f.sub.3 k.tau..sub.0) which is the negative of the center frequency of filter element f.sub.3. This has the effect of converting the signal output of filter element f.sub.3 to baseband. The output signals of each of the other filter elements of filter bank 216 (except filter element f.sub.1) are similarly processed, with the result that all the filter element output signals are converted to baseband signals with a bandwidth corresponding to that of the filter element. The bandwidth of each filter element of filter bank 216 is small, on the order of a few Hertz or less.
As described in the Urkowitz '706 patent, the pulse compression and range sidelobe suppression performed in processor 228 of FIG. 2 may be performed by a pair of FIR filters implemented as tapped delay lines with weighting and summing. The salient requirement is that the range sidelobe reduction function be provided individually for the signal component in each frequency bin. When this requirement is met, the range sidelobe suppression can be optimized for each frequency increment, and the suppression can be maintained. The combination 228 of pulse compressor and range sidelobe suppression follows each of the complex multipliers 220. Since each complex multiplication removes the residual Doppler phase shift across the uncompressed pulse, no residual Doppler phase shift remains on the uncompressed pulse. Each pulse compressor and range sidelobe suppressor is a zero Doppler design. All of the pulse compressor and range sidelobe suppressors are therefore identical, which is a cost advantage.
It was believed that the pulse-to-pulse Doppler filter bank, range mixers, pulse compressors, and sidelobe suppressors constituted a combination of time-invariant and time-variant filters, in which the order of processing is critical.